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19-0709; Rev 0; 3/07
KIT ATION EVALU ILABLE AVA
Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp
General Description
The MAX17010 contains a high-performance step-up switching regulator, a high-speed operational amplifier (op amp), and a high-voltage level-shifting scan driver. The device is optimized for thin-film transistor (TFT) liquidcrystal display (LCD) applications. The step-up DC-DC converter provides the regulated supply voltage for the panel-source driver ICs. The converter is a 1.2MHz current-mode regulator with an integrated 20V n-channel power MOSFET. The high switching frequency allows the use of ultra-small inductors and ceramic capacitors. The current-mode control architecture provides fast transient response to pulsed loads. The step-up regulator features undervoltage lockout (UVLO), soft-start, and internal current limit. The high-current op amp is designed to drive the LCD backplane (VCOM). The amplifier features high output current (150mA), fast slew rate (45V/s), wide bandwidth (20MHz), and rail-to-rail inputs and outputs. The high-voltage, level-shifting scan driver is designed to work with panels that incorporate row drivers on the panel glass. Its eight outputs swing from +30V (max) to -10V and can swiftly drive capacitive loads. The MAX17010 is available in a 40-pin thin QFN package with a maximum thickness of 0.8mm for ultra-thin LCD panels. The device operates over the -40C to +85C temperature range. Applications . Notebook Computer Displays LCD Monitor Panels
COMP SUP
Features
o 1.8 V to 5.5V IN Supply Voltage Range o 3mA SUP Quiescent Current (Switching) o 1.2MHz Current-Mode Step-Up Regulator Fast Transient Response High-Accuracy Output Voltage (1.0%) Built-In 20V, 1.9A, 200m MOSFET High Efficiency (> 85%) Digital Soft-Start o High-Speed Op Amp 150mA Output Current 45V/s Slew Rate 20MHz, -3dB Bandwidth o High-Voltage Level-Shifting Scan Drivers Logic-Level Inputs +30V to -10V Output Rails o Thermal-Overload Protection o 40-Pin, 5mm x 5mm, Thin QFN Package
MAX17010
Minimal Operating Circuit
VIN VMAIN
LX SHDN IN PGND FB
Ordering Information
PART MAX17010ETL+ TEMP RANGE -40C to +85C PIN-PACKAGE 40 Thin QFN-EP* (5mm x 5mm)
AGND VL POS
MAX17010
NEG GON1 GON2 A1 A2 A3 A4 A5 A6 A7 A8 VCOM BGND Y1 Y2 Y3 Y4 Y5 Y6 Y7 Y8 TO VCOM BACKPLANE
+Denotes a lead-free package. *EP = Exposed paddle.
EP
Pin Configuration appears at end of data sheet.
________________________________________________________________ Maxim Integrated Products 1
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim's website at www.maxim-ic.com.
Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp MAX17010
ABSOLUTE MAXIMUM RATINGS
IN, SHDN to GND ..................................................-0.3V to +7.5V VL to AGND ...........................................................-0.3V to +6.0V COMP, FB to GND ........................................-0.3V to (VL + 0.3V) VCOM, NEG, POS to BGND .....................-0.3V to (VSUP + 0.3V) LX to GND ..............................................................-0.3V to +20V SUP to GND............................................................-0.3V to +20V A_ to AGND ............................................................-0.3V to +20V A_ Input Current..................................................................20mA PGND, BGND to AGND.........................................-0.3V to +0.3V GON1, GON2 to AGND..........................................-0.3V to +32V GOFF to AGND......................................................-12V to + 0.3V Y1-Y6 to AGND.......................(VGOFF - 0.3V) to (VGON1 + 0.3V) Y7, Y8 to AGND.......................(VGOFF - 0.3V) to (VGON2 + 0.3V) LX, PGND RMS Current Rating.............................................2.4A Continuous Power Dissipation (TA = +70C) NiPd Lead Frame with Nonconductive Epoxy 40-Pin, 5mm x 5mm, Thin QFN (derate 35.7mW/C above +70C)........................................................................2857mW Operating Temperature Range ...........................-40C to +85C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = V SHDN = +3V, Circuit of Figure 1, SUP = 8.5V, VGON1 = VGON2 = 30V, VGOFF = -10V, VPOS = VNEG = 4V, TA = 0C to +85C. Typical values are at TA = +25C, unless otherwise noted.)
PARAMETER IN Input-Voltage Range IN Quiescent Current IN Undervoltage Lockout Thermal Shutdown VL Output Voltage VL Undervoltage Lockout VL Maximum Output Current MAIN DC-DC CONVERTER SUP Supply Current Operating Frequency Oscillator Maximum Duty Cycle FB Regulation Voltage FB Load Regulation FB Line Regulation FB Input Bias Current FB Transconductance FB Voltage Gain FB Fault-Timer Trip Threshold FB Undervoltage Switching Inhibit LX On-Resistance LX Leakage Current LX Current Limit Current-Sense Transresistance Soft-Start Period ILX = 200mA VLX = 13V 65% duty cycle 1.6 0.25 FB = COMP 0 < IMAIN < 200mA, transient only VIN = 1.8V to 5.5V VFB = 1.3V I = 5A at COMP FB to COMP Falling edge 0.96 50 50 75 VFB = 1.5V, no load VFB = 1.1V, no load 990 88 1.222 1.5 3.5 1170 92 1.235 -1 0 125 160 2400 1.00 100 200 0.01 1.9 0.42 3 1.04 150 330 20 2.2 0.55 200 280 2.5 4.5 1350 96 1.248 mA kHz % V % %/V nA S V/V V mV m A A V/A ms VL rising, 200mV hysteresis (typ) VFB = 1V VIN = 3V, VFB = 1.5V, not switching IN rising; typical hysteresis 100mV; LX remains off below this level Rising edge, 15oC hysteresis 3.8 2.4 10 CONDITIONS MIN 1.8 0.05 1.30 160 4.0 2.7 4.2 3.0 TYP MAX 5.5 0.10 1.75 UNITS V mA V
o
C
BOOTSTRAP LINEAR REGULATOR (VL) V V mA
2
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Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp
ELECTRICAL CHARACTERISTICS (continued)
(VIN = V SHDN = +3V, Circuit of Figure 1, SUP = 8.5V, VGON1 = VGON2 = 30V, VGOFF = -10V, VPOS = VNEG = 4V, TA = 0C to +85C. Typical values are at TA = +25C, unless otherwise noted.)
PARAMETER CONTROL INPUTS SHDN Input-Low Voltage SHDN Input-High Voltage Maximum SHDN Input Current OP AMP SUP Supply Range SUP Overvoltage Threshold SUP Undervoltage Threshold Input Offset Voltage Input Bias Current Input Common-Mode Voltage Range VCOM Output-Voltage Swing High IVCOM = 5mA (Note 1) (Note 2) VNEG, VPOS = VSUP / 2 VNEG, VPOS = VSUP / 2 -50 0 VSUP - 100 VSUP - 50 50 +75 -75 40 20 Short to VSUP / 2, sourcing Short to VSUP / 2, sinking 50 50 12 12 -10 A1-A8 = AGND, no load A1-A8 = AGND, no load A1-A8 = AGND, no load IOUT =10mA IOUT =10mA IOUT =10mA CLOAD = 100pF (Note 3) CLOAD = 100pF (Note 3) CLOAD = 100pF (Note 3) CLOAD = 100pF (Note 3) 50 VGON1 - 1.0 VGON2 - 1.0 75 30 10 VGOFF + 0.3 VGON1 - 0.3 VGON2 - 0.3 40 16 16 80 35 35 150 150 30 30 -5 125 60 20 VGOFF + 1.0 100 5 18.1 19.0 18 19.9 1.4 12 +50 VSUP V V V mV nA V mV mV mA mA V/s MHz mA 1.8V VIN 3.0V 3.0V VIN 5.5V 1.8 2.0 -1 +1 0.6 V V A CONDITIONS MIN TYP MAX UNITS
MAX17010
VCOM Output-Voltage Swing Low IVCOM = -5mA VCOM Output Current High VCOM Output Current Low Slew Rate -3dB Bandwidth VCOM Short-Circuit Current HIGH-VOLTAGE SCAN DRIVER GON1 Input-Voltage Range GON2 Input-Voltage Range GOFF Input-Voltage Range GOFF Supply Current GON1 Supply Current GON2 Supply Current Output-Voltage Low (Y1-Y8) Output-Voltage High (Y1-Y6) Output-Voltage High (Y7-Y8) Propagation Delay Rise Time (Y1-Y8) Fall Time (Y1-Y8) Maximum Operating Frequency VVCOM = VSUP - 1V VVCOM = 1V
V V V A A A V V V ns ns ns kHz
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Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp MAX17010
ELECTRICAL CHARACTERISTICS (continued)
(VIN = V SHDN = +3V, Circuit of Figure 1, SUP = 8.5V, VGON1 = VGON2 = 30V, VGOFF = -10V, VPOS = VNEG = 4V, TA = 0C to +85C. Typical values are at TA = +25C, unless otherwise noted.)
PARAMETER CONTROL INPUTS Logic Input-Voltage Threshold Rising (A1-A8) Logic Input-Voltage Threshold Falling (A1-A8) Logic Input-Voltage Hysteresis Logic Input Bias Current (A1-A8) VA1-A8 = 18V 1.2 0.7 1.6 0.9 0.7 20 45 2.0 1.12 V V V A CONDITIONS MIN TYP MAX UNITS
ELECTRICAL CHARACTERISTICS
(VIN = V SHDN = +3V, Circuit of Figure 1, SUP = 8V, VGON1 = VGON2 = 30, VGOFF = -10V, VPOS = VNEG = 4V, OE = 0V, TA = -40C to +85C.) (Note 4)
PARAMETER IN Input-Voltage Range IN Quiescent Current VIN = 3V, VFB = 1.5V, not switching IN rising; 100mV hysteresis (typ); LX remains off below IN Undervoltage Lockout this level BOOTSTRAP LINEAR REGULATOR (VL) VL Output Voltage VL Undervoltage Lockout VL Maximum Output Current MAIN DC-DC CONVERTER SUP Supply Current Operating Frequency Oscillator Maximum Duty Cycle FB Regulation Voltage FB Transconductance FB Fault Timer Trip Threshold FB Undervoltage Switching Inhibit LX On-Resistance LX Current Limit OP AMP SUP Supply Range SUP Overvoltage Fault Threshold SUP Undervoltage Fault Threshold Input Offset Voltage Input Common-Mode Voltage Range VCOM Output-Voltage Swing High VCOM Output-Voltage Swing Low VCOM Short-Circuit Current IVCOM = 5mA IVCOM = -5mA Short to VSUP / 2, sourcing Short to VSUP / 2 , sinking 50 50 (Note 1) (Note 2) VNEG, VPOS = VSUP / 2 0 VSUP - 100 100 5 18 18 19.9 1.4 12 VSUP V V V mV V mV mV mA ILX = 200mA 65% duty cycle 1.6 FB = COMP I = 5A at COMP Falling edge VFB = 1.5V, no load VFB = 1.1V, no load 990 88 1.216 75 0.96 50 2.8 5.0 1350 96 1.254 280 1.04 150 330 2.2 mA kHz % V S V mV m A VL rising, 200mV hysteresis (typ) VFB = 1V 3.8 2.4 10 CONDITIONS MIN 1.8 TYP MAX 5.5 0.1 1.75 UNITS V mA V
4.2 3.0
V V mA
4
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Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp
ELECTRICAL CHARACTERISTICS (continued)
(VIN = V SHDN = +3V, Circuit of Figure 1, SUP = 8V, VGON1 = VGON2 = 30, VGOFF = -10V, VPOS = VNEG = 4V, OE = 0V, TA = -40C to +85C.) (Note 4)
PARAMETER HIGH-VOLTAGE SCAN DRIVER GON1 Input-Voltage Range GON2 Input-Voltage Range GOFF Input-Voltage Range GOFF Supply Current GON1 Supply Current GON2 Supply Current Output-Voltage Low (Y1-Y8) Output-Voltage High (Y1-Y6) Output-Voltage High (Y7-Y8) CONTROL INPUTS Logic Input-Voltage Threshold Rising (A1-A8) Logic Input-Voltage Threshold Falling (A1-A8) Logic Input Bias Current (A1-A8) VA1-A8 = 18V 1.2 0.67 2.0 1.12 55 V V A A1-A8 = AGND, no load A1-A8 = AGND, no load A1-A8 = AGND, no load IOUT =10mA IOUT =10mA IOUT =10mA VGON1 -1 VGON2 -1 12 12 -10 30 30 -5 125 60 20 VGOFF +1 V V V A A A V V V CONDITIONS MIN TYP MAX UNITS
MAX17010
Note 1: Inhibits boost switching if SUP exceeds the overvoltage threshold. Switching resumes when SUP drops below the threshold. Note 2: Boost switching is not enabled until SUP is above undervoltage threshold. Note 3: Guaranteed by design, not production tested. Note 4: -40C specifications are guaranteed by design, not production tested.
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Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp MAX17010
Typical Operating Characteristics
(Circuit of Figure 1, VIN = 3V, VMAIN = 8.5V, TA = +25C, unless otherwise noted.)
STEP-UP CONVERTER LOAD REGULATION
MAX17010 toc02
STEP-UP CONVERTER EFFICIENCY
MAX17010 toc01
VL LOAD REGULATION
0 -0.05 -0.10 OUTPUT ERROR (%) OUTPUT ERROR (%) -0.15 -0.20 -0.25 -0.30 -0.35 -0.40 VIN = 3.3V 0.01 0.1 1 10 -0.8 -1.0 1 -0.2 -0.4 -0.6 0.2
90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 1
VIN = 5.0V VIN = 3.3V
VIN = 5.0V 0
VIN = 3.3V VIN = 1.8V
VIN = 1.8V
10
100
1000
10
100 LOAD CURRENT (mA)
1000
10,000
LOAD CURRENT (mA)
LOAD CURRENT (mA)
STEP-UP CONVERTER LINE REGULATION UNDER DIFFERENT LOADS
MAX17010 toc04
IN SUPPLY QUIESCENT CURRENT vs. IN SUPPLY VOLTAGE
MAX17010 toc05
INPUT SUPPLY CURRENT vs. TEMPERATURE
VIN = 5V 60 IN SUPPLY CURRENT (A) 50 40 30 20 10 0 NO LOAD ON VMAIN -60 -40 -20 0 20 40 60 80 100 TEMPERATURE (C) VIN = 3.3V
MAX17010 toc06
80 70 SUPPLY CURRENT (A) 60 50 40 30 20 10 0 0.2A LOAD NO LOAD
70
0.4 0.2 OUTPUT ERROR (%) 0 -0.2 -0.4 0.3A LOAD -0.6 0.2A LOAD -0.8 -1.0 1.8 2.3 2.8 3.3 3.8 4.3 4.8 5.3 5.8 INPUT VOLTAGE (V) 0.1A LOAD NO LOAD
1.6 2.1
2.6 3.1
3.6 4.1 4.6 5.1 5.6
SUPPLY VOLTAGE (V)
STEP-UP CONVERTER SWITCHING FREQUENCY vs. INPUT VOLTAGE
1.19 SWITCHING FREQUENCY (MHz) 1.18 1.17 1.16 1.15 1.14 1.13 1.12 1.11 1.10 1.6 2.1 2.6 3.1 3.6 4.1 4.6 5.1 5.6 INPUT VOLTAGE (V) 0mA 0V 0V 100mA LOAD
MAX17010 toc07
STEP-UP CONVERTER SOFT-START WITH HEAVY LOAD
MAX17010 toc08
1.20
LX 5V/div 0V
VMAIN 5V/div IL 500mA/div
SHDN CONTROL 5V/div 2ms/div
6
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MAX17010 toc03
100
Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp
Typical Operating Characteristics (continued)
(Circuit of Figure 1, VIN = 3V, VMAIN = 8.5V, TA = +25C, unless otherwise noted.)
STEP-UP CONVERTER LOAD-TRANSIENT RESPONSE (30mA TO 300mA)
MAX17010 toc09
MAX17010
STEP-UP CONVERTER PULSED LOADTRANSIENT RESPONSE (30mA TO 1A)
MAX17010 toc10
0V
VLX 10V/div IL 1A/div
0V
VLX 10V/div IL 1A/div VMAIN AC-COUPLED 200mV/div LOAD CURRENT 1A/div 10s/div
0A VMAIN AC-COUPLED 200mV/div LOAD CURRENT 200mA/div 0mA 100s/div
0A
0A
STEP-UP CONVERTER TIMER DELAY LATCH RESPONSE TO OVERLOAD
MAX17010 toc11
POWER-UP SEQUENCE OF ALL SUPPLY OUTPUTS
MAX17010 toc12
0V
VLX 10V/div VMAIN 5V/div
0V
VL 5V/div VMAIN 5V/div VGON 20V/div VCOM 5V/div VIN 5V/div VGOFF 10V/div SHDN CONTROL 5V/div
0V 0V
0V IL 2A/div 0A LOAD CURRENT 1A/div 10ms/div
0V 0V
0A
0V 2ms/div
SUP SUPPLY CURRENT vs. TEMPERATURE
MAX17010 toc13
OPERATIONAL AMPLIFIER FREQUENCY RESPONSE
MAX17010 toc14
OPERATIONAL AMPLIFIER POWER-SUPPLY REJECTION RATIO
MAX17010 toc15
3.0 VIN = 3.3V SUP SUPPLY CURRENT (mA) 2.5 2.0
10 5 NO LOAD 0
0 -10 -20 PSRR (dB) -30 -40
1.5 1.0 0.5 NO LOAD ON VMAIN 0 -60 -40 -20 0 20 40 60 80 100 TEMPERATURE (C)
GAIN (dB)
VIN = 5.0V
-5 -10 100pF LOAD -15 -20 AV = 1V VIN = 3.3V 100 1k 10k 100k
-50 -60 10 100 1k FREQUENCY (Hz) 10k 100k FREQUENCY (Hz)
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Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp MAX17010
Typical Operating Characteristics (continued)
(Circuit of Figure 1, VIN = 3V, VMAIN = 8.5V, TA = +25C, unless otherwise noted.)
OPERATIONAL AMPLIFIER RAIL-TO-RAIL INPUT/OUTPUT WAVEFORMS
MAX17010 toc16
OPERATIONAL AMPLIFIER LOAD-TRANSIENT RESPONSE
MAX17010 toc17
OPERATIONAL AMPLIFIER LARGE-SIGNAL STEP RESPONSE
MAX17010 toc18
VVCOM VPOS 5V/div 0V
(AC-COUPLED)
0mV
100mV/div 0V
VPOS 5V/div
VVCOM 5V/div 0V 10s/div
IVCOM 50mA/div 0mA
0V
VVCOM 5V/div
20s/div
40s/div
OPERATIONAL AMPLIFIER SMALL-SIGNAL STEP RESPONSE
MAX17010 toc19
SCAN DRIVER INPUT/OUTPUT WAVEFORMS WITH LOGIC INPUT
MAX17010 toc20
VPOS (AC-COUPLED) 100mV/div
0V
VA 5V/div
VVCOM (AC-COUPLED) 100mV/div
VY 10V/div 0V
40s/div
4s/div
SCAN DRIVER PROPAGATION DELAY (RISING EDGE)
MAX17010 toc21
SCAN DRIVER PROPAGATION DELAY (FALLING EDGE)
MAX17010 toc22
0V
VA 5V/div
0V
VA 5V/div
VY 10V/div 0V 0V
VY 10V/div
100ns/div
100ns/div
8
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Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp
Pin Description
PIN 1, 24, 30, 31, 40 2, 3 4 5 6 7 8-11 12-19 20-23 25 26 27 28 29 32 33 34 35 36 37 38, 39 -- NAME N.C. PGND FB AGND GON1 GOFF A1-A4 Y1-Y8 A5-A8 GON2 AGND COMP VL BGND SUP POS NEG VCOM SHDN IN LX EP No Connection. Not internally connected. Power Ground. Source connection of the internal step-up regulator power switch. Feedback Pin. Connect external resistor-divider tap here and minimize trace area. Set VOUT according to: VOUT = 1.235V (1 + R1/R2) (Figure 1). Ground Gate-On Supply. GON1 is the positive supply for the Y1-Y6 level-shifter circuitry. Bypass to AGND with a minimum 0.1F ceramic capacitor. Gate-Off Supply. GOFF is the negative supply voltage for the Y1-Y8 high-voltage driver outputs. Bypass to AGND with a minimum 0.1F ceramic capacitor. High-Voltage-Driver Logic-Level Inputs Level-Shifter High-Voltage Outputs High-Voltage-Driver Logic-Level Inputs Gate-On Supply. GON2 is the positive supply for the Y7 and Y8 level-shifter circuitry. Bypass to AGND with a minimum 0.1F ceramic capacitor. Ground. Internally connected to pin 5. Compensation Pin for Error Amplifier. Connect a series RC from this pin to AGND. Typical values are 100k and 220pF. 4V On-Chip Regulator Output. This regulator powers internal analog circuitry for the boost and op amp. Bypass VL to AGND with a 0.22F or greater ceramic capacitor. Amplifier Ground Op Amp and Internal VL Linear Regulator Supply Input. Bypass SUP to BGND with a 0.1F capacitor. Op Amp Noninverting Input Op Amp Inverting Input Op Amp Output Shutdown Control Input. Pull SHDN low to turn off the DC-DC converter and high-voltage drivers only (VL and op amp remain on). Supply Pin. Bypass to AGND with a minimum 0.1F ceramic capacitor. Switching Node. Connect inductor/catch diode here and minimize trace area for lowest EMI. Exposed Backside Paddle FUNCTION
MAX17010
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Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp MAX17010
VGON 0.1F VGOFF 0.1F VIN +2.7V TO +5.5V C1 10F 6.3V L1 3.6H 0 SHDN IN 1F PGND COMP CCOMP 220pF RCOMP 100k AGND VL 0.22F VGON VGOFF POS R6 200k TO VCOM BACKPLANE LX FB R2 34k 1% D1 R1 200k 1% 0.1F 0.1F D3 D4 0.1F VMAIN +8.5V/300mA C2 4.7F 10V C3 4.7F 10V 0.1F
D2
SUP 0.1F R5 200k
MAX17010
NEG GON1 GON2 GOFF A1 A2 A3 A4 A5 A6 A7 A8 EP VCOM BGND Y1 Y2 Y3 Y4 Y5 Y6 Y7 Y8
Figure 1. MAX17010 Typical Application Circuit
Typical Application Circuit
The MAX17010 typical application circuit (Figure 1) generates a +8.5V source-driver supply and approximately +22V and -7V gate-driver supplies for TFT displays. The input voltage range for the IC is from +1.8V to +5.5V, but the Figure 1 circuit is designed to run from 2.7V to 5.5V. Table 1 lists the recommended components and Table 2 lists the contact information of component suppliers.
Table 1. Component List
DESIGNATION C1 C2, C3 D1 D2, D3, D4 L1 DESCRIPTION 10F, 6.3V X5R ceramic capacitor (1206) TDK C3216X5ROJ106M 4.7F, 10V X5R ceramic capacitors (1206) TDK C3216X5R1A475M 3A, 30V Schottky diode (M-flat) Toshiba CMS02 200mA, 100V, dual, ultra-fast diodes (SOT23) Fairchild MMBD4148SE 3.6H, 1.8A inductor Sumida CMD6D11BHPNP-3R6MC
10
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Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp
Table 2. Component Suppliers
SUPPLIER Fairchild Sumida TDK PHONE 408-822-2000 847-545-6700 847-803-6100 FAX 408-822-2102 847-545-6720 847-390-4405 WEBSITE www.fairchildsemi.com www.sumida.com www.component.tdk.com www.toshiba.com/taec
MAX17010
Toshiba 949-455-2000 949-859-3963 Note: Indicate that you are using the MAX17010 when contacting these component suppliers.
VIN IN VL LINEAR REGULATOR AND BOOTSTRAP SHDN
L
D
VMAIN
LX
Y1-Y6
STEP-UP REGULATOR CONTROLLER
PGND FB
GON1 COMP AGND
A1-A6 GON2 A7, A8 +
SUP NEG VCOM TO VCOM BACKPLANE
GOFF Y7, Y8 POS
MAX17010
BGND
Figure 2. MAX17010 Functional Diagram
Detailed Description
The MAX17010 contains a high-performance step-up switching regulator, a high-speed op amp, and a highvoltage, level-shifting scan driver optimized for activematrix TFT LCDs. Figure 2 shows the MAX17010 functional diagram.
Step-Up Regulator
The step-up regulator employs a current-mode, fixed-frequency PWM architecture to maximize loop bandwidth
and provide fast transient response to pulsed loads found in source drivers of TFT LCD panels. The high switching frequency (1.2MHz) allows the use of low-profile inductors and ceramic capacitors to minimize the thickness of LCD panel designs. The integrated high-efficiency MOSFET and the IC's built-in digital soft-start functions reduce the number of external components required while controlling inrush current. The output voltage can be set from 5V to 18V with an external resistive voltage-divider.
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Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp MAX17010
The regulator controls the output voltage, and the power delivered to the output, by modulating the duty cycle (D) of the internal power MOSFET in each switching cycle. The duty cycle of the MOSFET is approximated by: V -V D MAIN IN VMAIN Figure 3 shows the block diagram of the step-up regulator. An error amplifier compares the signal at FB to 1.235V and changes the COMP output. The voltage at COMP determines the current trip point each time the internal MOSFET turns on. As the load varies, the error amplifier sources or sinks current to the COMP output accordingly, to produce the inductor peak current necessary to service the load. To maintain stability at high duty cycles, a slope-compensation signal is summed with the current-sense signal. On the rising edge of the internal clock, the controller sets a flip-flop, turning on the n-channel MOSFET, and applying the input voltage across the inductor. The current through the inductor ramps up linearly, storing energy in its magnetic field. Once the sum of the current-feedback signal and the slope compensation exceed the COMP voltage, the controller resets the flipflop and turns off the MOSFET. Since the inductor current is continuous, a transverse potential develops across the inductor that turns on the diode (D1). The voltage across the inductor then becomes the difference between the output voltage and the input voltage. This discharge condition forces the current through the inductor to ramp back down, transferring the energy stored in the magnetic field to the output capacitor and the load. The MOSFET remains off for the rest of the clock cycle.
Undervoltage Lockout (UVLO) The undervoltage lockout (UVLO) circuit compares the input voltage at IN with the UVLO threshold (1.3V rising and 1.2V falling) to ensure that the input voltage is high enough for reliable operation. The 100mV (typ) hysteresis prevents supply transients from causing a restart. Once the input voltage exceeds the UVLO rising threshold, startup begins. When the input voltage falls below the UVLO falling threshold, the controller turns off the main step-up regulator and the linear regulator outputs, disables the switch-control block, and the op amp outputs are high impedance.
CLOCK LOGIC AND DRIVER
LX
PGND CURRENT-LIMIT COMPARATOR + SLOPE COMP PWM COMPARATOR 1.2MHz OSCILLATOR SOFTSTART ILIMIT
+ -
CURRENT SENSE
TO FAULT LOGIC + FAULT COMPARATOR 1.0V
ERROR AMP + 1.235V COMP FB
Figure 3. Step-Up Regulator Block Diagram
12 ______________________________________________________________________________________
Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp
Linear Regulator (VL) The MAX17010 includes an internal 4V linear regulator. SUP is the input of the linear regulator. The input voltage range is between 5V and 18V. The output of the linear regulator (VL) is set to 4V (typ). The regulator powers all the internal circuitry including the MOSFET gate driver. Bypass the VL pin to AGND with a 0.22F or greater ceramic capacitor. SUP should be directly connected to the output of the step-up regulator. This feature significantly improves the efficiency at low input voltages. Bootstrapping and Soft-Start The MAX17010 features bootstrapping operation. In normal operation, the internal linear regulator supplies power to the internal circuitry. The input of the linear regulator (SUP) should be directly connected to the output of the step-up regulator. The MAX17010 is enabled when the input voltage at SUP is above 1.4V and the fault latch is not set. After being enabled, the regulator starts openloop switching to generate the supply voltage for the linear regulator. Step-up switching is inhibited if the stepup output voltage (VMAIN) exceeds the voltage on the SUP input. The internal reference block turns on when the VL voltage exceeds 2.7V (typ). When the reference voltage reaches regulation, the PWM controller and the current-limit circuit are enabled and the step-up regulator enters soft-start. During soft-start, the main step-up regulator directly limits the peak inductor current, allowing from zero up to the full current-limit value in 128 equal current steps. The maximum load current is available after the output voltage reaches regulation (which terminates soft-start), or after the soft-start timer expires in approximately 3ms. The soft-start routine minimizes the inrush current and voltage overshoot, and ensures a well-defined startup behavior. Fault Protection During steady-state operation, the MAX17010 monitors the FB voltage. If the FB voltage does not exceed 1V (typ), the MAX17010 activates an internal fault timer. If there is a continuous fault for the fault-timer duration, the MAX17010 sets the fault latch, shutting down all the outputs except VL. Once the fault condition is removed, cycle the input voltage to clear the fault latch and reactivate the device. The fault-detection circuit is disabled during the soft-start time. The MAX17010 monitors the SUP voltage for undervoltage and overvoltage conditions. If the SUP voltage is below 1.4V (typ) or above 19V (typ), the MAX17010 disables the gate driver of the step-up regulator and prevents the internal MOSFET from switching. The SUP undervoltage and overvoltage conditions do not set the fault latch.
Op Amps
The MAX17010 has an op amp that is typically used to drive the LCD backplane (VCOM) and/or the gammacorrection-divider string. The op amp features 150mA output short-circuit current, 45V/s slew rate, and 12MHz bandwidth. While the op amp is a rail-to-rail input and output design, its accuracy is significantly degraded for input voltages within 1V of its supply rails (SUP and VGND).
MAX17010
Short-Circuit Current Limit The op amp limits short-circuit current to approximately 150mA if the output is directly shorted to SUP or to AGND. If the short-circuit condition persists, the junction temperature of the IC rises until it reaches the thermalshutdown threshold (+160C typ). Once the junction temperature reaches the thermal-shutdown threshold, an internal thermal sensor immediately sets the thermal fault latch, shutting off all the IC's outputs except VL. The device remains inactive until the input voltage is cycled. Driving Pure Capacitive Load The op amp is typically used to drive the LCD backplane (VCOM) or the gamma-correction-divider string. The LCD backplane consists of a distributed series capacitance and resistance, a load that can be easily driven by the op amp. However, if the op amp is used in an application with a pure capacitive load, steps must be taken to ensure stable operation. As the op amp's capacitive load increases, the amplifier's bandwidth decreases and the gain peaking increases. A 5 to 50 small resistor placed between VCOM and the capacitive load reduces peaking but also reduces the gain. An alternative method of reducing peaking is to place a series RC network (snubber) in parallel with the capacitive load. The RC network does not continuously load the output or reduce the gain.
High-Voltage Level-Shifting Scan Driver
The MAX17010 includes eight logic-level to high-voltage level-shifting buffers, which can buffer eight logic inputs (A1-A8) and shift them to a desired level (Y1-Y8) to drive TFT-LCD row logic. The driver outputs, Y1-Y8, swing between their power-supply rails, according to the input-logic level on A1-A8. The driver output is GOFF when its respective input is logic low, and GON_ when its respective input is logic high. These eight driver channels are grouped for different high-level supplies. A1-A6 are supplied from GON1, and A7 and A8 are supplied from GON2. GON1 and GON2 can be tied together to make A1-A8 use identical supplies.
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13
Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp MAX17010
The high-voltage, level-shifting scan drivers are designed to drive the TFT panels with row-drivers integrated on the panel glass. Its eight outputs swing from +30V (max) to -6.3V (min) and can swiftly drive capacitive loads. The typical propagation delays are 40ns, with fast 16ns rise-and-fall times. The buffers can operate at frequencies up to 50kHz. and ratio of inductor resistance to other power-path resistances, the best LIR can shift up or down. If the inductor resistance is relatively high, more ripple can be accepted to reduce the number of turns required and increase the wire diameter. If the inductor resistance is relatively low, increasing inductance to lower the peak current can decrease losses throughout the power path. If extremely thin high-resistance inductors are used, as is common for LCD panel applications, the best LIR can increase to between 0.5 and 1.0. Once a physical inductor is chosen, higher and lower values of the inductor should be evaluated for efficiency improvements in typical operating regions. Calculate the approximate inductor value using the typical input voltage (VIN), the maximum output current (IMAIN(MAX)), the expected efficiency (TYP) taken from an appropriate curve in the Typical Operating Characteristics, and an estimate of LIR based on the above discussion:
2 TYP V VMAIN - VIN L = IN VMAIN IMAIN(MAX) x fOSC LIR
Thermal-Overload Protection
The thermal-overload protection prevents excessive power dissipation from overheating the device. When the junction temperature exceeds TJ = +160C, a thermal sensor immediately activates the fault protection, which shuts down all outputs except VL, allowing the device to cool down. Once the device cools down by approximately 15C, cycle the input voltage (below the UVLO-falling threshold) to clear the fault latch and reactivate the device. The thermal-overload protection protects the controller in the event of fault conditions. For continuous operation, do not exceed the absolute maximum junction temperature rating of TJ = +150C.
Design Procedure
Main Step-Up Regulator
Inductor Selection The minimum inductance value, peak current rating, and series resistance are factors to consider when selecting the inductor. These factors influence the converter's efficiency, maximum output-load capability, transient response time, and output-voltage ripple. Physical size and cost are also important factors to be considered. The maximum output current, input voltage, output voltage, and switching frequency determine the inductor value. Very high inductance values minimize the current ripple and therefore reduce the peak current, which decreases core losses in the inductor and I2R losses in the entire power path. However, large inductor values also require more energy storage and more turns of wire, which increase physical size and can increase I2R losses in the inductor. Low inductance values decrease the physical size but increase the current ripple and peak current. Finding the best inductor involves choosing the best compromise between circuit efficiency, inductor size, and cost. The equations used here include a constant (LIR), which is the ratio of the inductor peak-to-peak ripple current to the average DC inductor current at the fullload current. The best trade-off between inductor size and circuit efficiency for step-up regulators generally has an LIR between 0.3 and 0.5. However, depending on the AC characteristics of the inductor core material
14
Choose an available inductor value from an appropriate inductor family. Calculate the maximum DC input current at the minimum input voltage VIN(MIN) using conservation of energy and the expected efficiency at that operating point (MIN) taken from an appropriate curve in the Typical Operating Characteristics: IIN(DCMAX) = , IMAIN(MAX) x VMAIN VIN(MIN) x MIN
Calculate the ripple current at that operating point and the peak current required for the inductor: VIN(MIN) x (VMAIN - VIN(MIN) ) IRIPPLE = L x VMAIN x fOSC I IPEAK = IIN(DCMAX) + RIPPLE , 2 The inductor's saturation current rating and the MAX17010's LX current limit (ILIM) should exceed IPEAK and the inductor's DC current rating should exceed IIN(DC,MAX). For good efficiency, choose an inductor with less than 0.1 series resistance. Considering the Typical Operating Circuit, the maximum load current (IMAIN(MAX)) is 300mA, with an 8.5V output and a typical input voltage of 3V. Choosing an LIR of 0.45 and estimating efficiency of 85% at this operating point: 3V 8.5V - 3V 0.85 L= 3.6H 8.5V 0.3A x 1.2MHz 0.5
2
______________________________________________________________________________________
Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp
Using the circuit's minimum input voltage (2.2V) and estimating efficiency of 80% at that operating point: IIN(DCMAX) = , 0.3A x 8.5V 1.45A 2.2V x 0.8
The ripple current and the peak current are: IRIPPLE = 3.6H x 8.5V x 1.2MHz 2.2V x (8.5V - 2.2V ) 0.38A
Rectifier Diode The MAX17010's high switching frequency demands a high-speed rectifier. Schottky diodes are recommended for most applications because of their fast recovery time and low forward voltage. In general, a 2A Schottky diode complements the internal MOSFET well. Output Voltage Selection The output voltage of the main step-up regulator is adjusted by connecting a resistive voltage-divider from the output (VMAIN) to AGND with the center tap connected to FB (see Figure 1). Select R2 in the 10k to 50k range. Calculate R1 with the following equation:
V R1 = R2 x MAIN - 1 VREF where VREF, the step-up regulator's feedback set point, is 1.235V. Place R1 and R2 close to the IC.
MAX17010
IPEAK = 1.45A +
0.38A 1.64A 2
Output Capacitor Selection The total output-voltage ripple has two components: the capacitive ripple caused by the charging and discharging of the output capacitance, and the ohmic ripple due to the capacitor's equivalent series resistance (ESR): VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR)
I V -V VRIPPLE(C) MAIN MAIN IN COUT VMAINfOSC and: VRIPPLE(ESR) IPEAKRESR(COUT) where I PEAK is the peak inductor current (see the Inductor Selection section). For ceramic capacitors, the output-voltage ripple is typically dominated by VRIPPLE(C). The voltage rating and temperature characteristics of the output capacitor must also be considered.
Loop Compensation Choose RCOMP to set the high-frequency integrator gain for fast transient response. Choose CCOMP to set the integrator zero to maintain loop stability. For low-ESR output capacitors, use the following equations to obtain stable performance and good transient response:
RCOMP 1000 x VIN x VOUT x COUT L x IMAIN(MAX) VOUT x COUT 10 x IMAIN(MAX) x RCOMP
CCOMP
Input Capacitor Selection The input capacitor (CIN) reduces the current peaks drawn from the input supply and reduces noise injection into the IC. A 10F ceramic capacitor is used in the Typical Applications Circuit (Figure 1) because of the high source impedance seen in typical lab setups. Actual applications usually have much lower source impedance since the step-up regulator often runs directly from the output of another regulated supply. Typically, CIN can be reduced below the values used in the Typical Applications Circuit. Ensure a low-noise supply at IN by using adequate C IN . Alternatively, greater voltage variation can be tolerated on CIN if IN is decoupled from C IN using an RC lowpass filter, as shown in Figure 1.
To further optimize transient response, vary RCOMP in 20% steps and CCOMP in 50% steps, while observing transient response waveforms.
Applications Information
Power Dissipation
An IC's maximum power dissipation depends on the thermal resistance from the die to the ambient environment, and the ambient temperature. The thermal resistance depends on the IC package, PCB copper area, other thermal mass, and airflow. The MAX17010, with its exposed backside paddle soldered to an internal ground layer in a typical multilayer PCB, can dissipate about 2.8W into +70C still air. More PCB copper, cooler ambient air, and more airflow increase the possible dissipation, while less copper or warmer air decreases the IC's dissipation capability. The major components of power dissipation are the power dissipated in the step-up regulator and the power dissipated by the op amps.
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______________________________________________________________________________________
Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp MAX17010
Step-Up Regulator The largest portions of power dissipation in the step-up regulator are the internal MOSFET, inductor, and the output diode. If the step-up regulator has 90% efficiency, about 3% to 5% of the power is lost in the internal MOSFET, about 3% to 4% in the inductor, and about 1% in the output diode. The remaining 1% to 3% is distributed among the input and output capacitors and the PCB traces. If the input power is about 5W, the power lost in the internal MOSFET is about 150mW to 250mW. Op Amp The power dissipated in the op amp depends on its output current, the output voltage, and the supply voltage:
PDSOURCE = IVCOM(SOURCE) x (VSUP - VVOUT ) PDSINK = IVCOM(SINK) x VVOUT where IVCOM(SOURCE) is the output current sourced by the op amp, and IVCOM(SINK) is the output current that the op amp sinks. In a typical case where the supply voltage is 8.5V, and the output voltage is 4V with an output source current of 30mA, the power dissipated is 135mV. 2) Create a power ground island (PGND) consisting of the input- and output-capacitor grounds, PGND pin, and any charge-pump components. Connect all these together with short, wide traces or a small ground plane. Maximizing the width of the powerground traces improves efficiency and reduces output-voltage ripple and noise spikes. Create an analog ground plane (AGND) consisting of the AGND pin, all the feedback-divider ground connections, the op-amp-divider ground connections, the COMP capacitor ground connection, the SUP and VL bypass-capacitor ground connections, and the device's exposed backside pad. Connect the AGND and PGND islands by connecting the PGND pin directly to the exposed backside pad. Make no other connections between these separate ground planes. 3) Place the feedback-voltage-divider resistors as close to the feedback pin as possible. The divider's center trace should be kept short. Placing the resistors far away causes the FB trace to become an antenna that can pick up switching noise. Care should be taken to avoid running the feedback trace near LX or the switching nodes in the charge pumps. 4) Place the IN pin and VL pin bypass capacitors as close to the device as possible. The ground connections of the IN and VL bypass capacitors should be connected directly to the AGND pin or the IC's backside pad with a wide trace. 5) Minimize the length and maximize the width of the traces between the output capacitors and the load for best transient responses. 6) Minimize the size of the LX node while keeping it wide and short. Keep the LX node away from the feedback node and analog ground. Use DC traces as shield if necessary. 7) Refer to the MAX17010 evaluation kit for an example of proper board layout.
PCB Layout and Grounding
Careful PCB layout is important for proper operation. Use the following guidelines for good PCB layout: 1) Minimize the area of high-current loops by placing the inductor, output diode, and output capacitors near the input capacitors and near the LX and PGND pins. The high-current input loop goes from the positive terminal of the input capacitor to the inductor, to the IC's LX pins, out of PGND, and to the input capacitor's negative terminal. The highcurrent output loop is from the positive terminal of the input capacitor to the inductor, to the output diode (D1), to the positive terminal of the output capacitors, reconnecting between the outputcapacitor and input-capacitor ground terminals. Connect these loop components with short, wide connections. Avoid using vias in the high-current paths. If vias are unavoidable, use many vias in parallel to reduce resistance and inductance.
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Internal-Switch Boost Regulator with Integrated High-Voltage Level Shifter and Op Amp
Pin Configuration
COMP BGND AGND GON2 N.C.
Chip Information
TRANSISTOR COUNT: 9202 PROCESS: BiCMOS
MAX17010
VL
A8
A7
30 29 28 27 26 25 24 23 22 21 N.C. 31 SUP 32 POS 33 NEG 34 VCOM 35 SHDN 36 IN 37 LX 38 LX 39 N.C. 40 1 N.C. 2 PGND 3 PGND 4 FB 5 AGND 6 GON1 7 GOFF 8 A1 9 A2 10 A3 20 A5 19 Y8 18 Y7 17 Y6 16 Y5
A6
TOP VIEW
N.C.
Package Information
For the latest package outline information, go to www.maxim-ic.com/packages.
MAX17010
15 Y4 14 Y3 13 Y2 12 Y1 11 A4
THIN QFN (5mm x 5mm)
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 17
(c) 2007 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.

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